High power RF switch with controlled well voltage for improved linearity

ABSTRACT

RF transistors manufactured using a bulk CMOS process exhibit non-linear drain-body and source-body capacitances which degrade the linearity performance of the RF circuits implementing such transistors. The disclosed methods and devices address this issue and provide solutions based on implementing two or more bias voltages in accordance with the states of the transistors. Various exemplary RF circuits benefiting from the described methods and devices are also presented.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. patent applicationSer. No. 17/129,568, filed Dec. 21, 2020, titled “High Power RF SwitchWith Controlled Well Voltage For Improved Linearity,” now U.S. Pat. No.11,290,105 issued Mar. 29, 2022, which is herein incorporated byreference in its entirety.

FIELD

The present disclosure is related to methods and apparatus usingcontrolled well voltages for improved linearity in RF circuits, more inparticular in RF circuits including high power RF switches in a bulkCMOS process.

BACKGROUND

High power radio frequency (RF) switches with high linearity aregenerally manufactured in silicon-on-insulator (SOI) processes byimplementing floating body devices. As known in the art, because of thereduction of drain-body or source-body capacitances, floating bodymetal-oxide-semiconductor field effect transistors (MOSFETs) provide abetter overall linearity compared, for example, with a bulk CMOSprocess. This is particularly important during operative conditions,when some switches are in an OFF state in which presence of undesiredjunction capacitances can degrade the overall linearity of the circuit.At the same time, a bulk CMOS process provides cost advantages and ishighly scalable. It is therefore desirable to be able to manufacturehigh linearity switches in a bulk CMOS process to benefit from a bettercost-performance tradeoff.

FIG. 1A shows a prior art transistor built in accordance with an SOIprocess. Device capacitances between terminals of the transistors, i.e.gate-source, gate-drain, and drain-source capacitances (Cgs, Cgd, Cds)are also shown by way of schematic capacitors. FIG. 1B on the otherhand, shows the same device manufactured according to a bulk CMOSprocess where, in addition to the mentioned gate-source and gate-draindevice capacitances, variable drain-body, variable source-body, andgate-body capacitances (Cdb, Csb, Cgb) are also present. FIG. 1C shows atop view layout of the transistor of FIG. 1B.

FIG. 2A shows a cross section of a prior art N-channel field effecttransistor (NFET) device built with triple-well bulk CMOS process. FIG.2B shows the electrical model of the NFET shown in FIG. 2A. As shown inFIGS. 2A-2B, the drain-body and source-body junctions are representedthrough respective drain-body and source-body diodes (Ddb, Dsb) havingnon-linear capacitances which degrade linearity when the NFET device ismodulated with an RF signal.

In view of the above, solutions are therefore needed to reduce theundesired effect of the junction capacitances of devices manufacturedusing a bulk CMOS process.

SUMMARY

The disclosed methods and devices provide practical solutions to theabove-mentioned problem.

According to a first aspect, a controllable field effect transistor(FET) for use in a radio frequency (RF) switch is disclosed, the FETcomprising gate, source, drain, body, substrate and a well configured toreceive a well bias voltage, wherein: the well is of an oppositesemiconductor polarity to the body and the substrate of the FET; thewell is disposed in a region separate from the body and from thesubstrate of the FET; the FET is fabricated with a bulk complementarymetal-oxide-semiconductor (CMOS) process and is configured to: receive agate bias voltage switchable between a first gate bias voltage level anda second gate bias voltage level to put the FET in an ON or OFF staterespectively, receive a body bias voltage switchable between a firstbody bias voltage level in correspondence with the ON state, and asecond body bias voltage level in correspondence with the OFF state, andreceive a well bias voltage switchable between a first well bias voltagelevel in correspondence with the ON state, and a second well biasvoltage level in correspondence with the OFF state.

According to a second aspect, a method of biasing a radio frequency (RF)field effect transistor (FET) switch manufactured using a bulk CMOSprocess is disclosed, the method comprising: in an ON state of the FETswitch: applying a first gate voltage to a gate terminal of the FETswitch, applying a first body voltage to a body terminal of the FETswitch, and applying a first well voltage to a well terminal of the FETswitch; in an OFF state of the FET switch: applying a second gatevoltage different from the first gate voltage to the gate terminal ofthe FET switch; applying a second body voltage different from the firstbody voltage to the body terminal of the FET switch, and applying asecond well voltage different from the first well voltage to the wellterminal of the FET switch.

Further aspects of the disclosure are provided in the description,drawings and claims of the present application.

DESCRIPTION OF THE DRAWINGS

FIG. 1A shows a prior art transistor built with a SOI process.

FIG. 1B shows a prior art transistor built with a bulk CMOS process.

FIG. 1C shows the layout of the prior art transistor of FIG. 1B

FIG. 2A shows a schematic cross section of a prior art NFET device builtwith a bulk CMOS process.

FIG. 2B shows an equivalent electrical model of the NFET shown in FIG.2A.

FIG. 3 shows an exemplary circuit (300) according to the teachings ofthe present disclosure.

FIGS. 4A-4C show exemplary RF switches adopting the teachings of thepresent disclosure.

FIG. 5 shows an exemplary RF switch according to an embodiment of thepresent disclosure.

DETAILED DESCRIPTION

Throughout the present disclosure, the term “stress voltages” in a FETrefers to the time-dependent dielectric breakdown voltage of the gate,and the gate oxide rupture voltage.

Referring back to FIG. 1B, junction capacitances can generally becalculated using the following formula:

$\begin{matrix}{C_{j} = \frac{C_{jo}}{\left( {1 + {V_{r}/\varnothing_{b}}} \right)^{m}}} & (1)\end{matrix}$wherein V_(r) is the absolute value of the reverse voltage (defined asthe voltage to keep the junction in a reverse biased condition, alsoknown as reverse bias) across the junction, Ø_(b) is the built-inpotential of the junction and m is a number typically within the rangeof 0.3 to 0.4. FIG. 1C shows the layout of the NFET of FIG. 1B, whereinthe width of the device is indicated with (W) and the width of thesource (S) and drain (D) is indicated with (EThe drain-body andsource-body capacitances (Cdb, Csb) can now be calculated as:C _(db) =C _(sb) =W*E*C _(j)+2*(W+E)*C _(jsw)  (2)wherein C_(j) is the bottom plate junction capacitance and C_(jsw) isthe junction sidewall capacitance. Looking at equations (1) and (2), theinventors have observed that one way to reduce the junction capacitancesin a bulk CMOS process is to increase the reverse bias across suchjunctions. As will be explained more in detail below, the disclosedmethods and devices leverage such observation to provide a better costperformance tradeoff when manufacturing RF circuits based on the bulkCMOS process.

FIG. 3 shows a circuit (300) in accordance with the teachings of thepresent disclosure showing biasing control of a device during operationto mitigate the undesired impact of non-linear junction capacitances.Circuit (300) comprises a transistor (T1) (e.g. an NFET), and a controlcircuit (310) receiving control signal (Vc) and including level shifters(L1, L2, L3). Each of level shifters (L1, L2) may be configured to haveDC supply voltages (V11, V12), and (V21 V22). Level shifter (L3) mayoperate with DC bias voltages (Vdd1, Vdd2), i.e. the biasing of theN-well (NW) of transistor (T1) can be switched between such DC biasvoltages. Level shifter (L1) is coupled to the gate terminal oftransistor (T1) through gate resistor (Rg), while level shifter (L2) iscoupled to the body terminal of transistor (T1) via body resistor(Rbody). The source terminal of transistor (T1) may be shorted to ground(not shown in the figure) in a shunt configuration and drain-sourceresistor (Rds) connects the drain and source terminals together.According to the teachings of the present disclosure, the source anddrain terminals may have a path to ground such that their DC voltagesare at ground potential.

With continued reference to FIG. 3 according to embodiments of thepresent disclosure, when transistor (T1) is enabled, the gate and bodyvoltages of transistor (T1) are set to voltages (V11, V21),respectively. As a result, a voltage difference of V=V11-V21 will appearacross the gate-body terminals of transistor (T1). On the other hand,when transistor (T1) is disabled, its gate voltage is set to voltage(V12) and its body voltage is set to (V22). In some embodiments, thesource terminal may be grounded (not shown in FIG. 3 ) and also coupledto the drain terminal of transistor (T1), both of the source and drainterminals will be sitting at 0V. As a result, when transistor (T1) isdisabled, i.e. the drain-body and the source-body junctions arereverse-biased and will have a voltage equal to DC voltage (V22).

In accordance with the teachings of the present disclosure, DC voltages(V11, V12, V21, V22) may be chosen such that in the disabled state,there is a reverse bias across the drain-body and the source-bodyjunctions of transistor (T1), without causing breakdown of thedrain-body and source-body junctions when in the disabled state andwithout overstressing the gate-body region of transistor (T1) when inthe enabled state. Moreover, DC voltage (Vdd2) may be smaller than DCvoltage (Vdd1). According to the teachings of the present disclosure, inthe disabled mode, the reverse bias voltages across source-body anddrain-body (Vsb and Vdb) are maximized while maintaining Vgb<Vtddb andVgb<Vox-rupture in enabled mode. Vgb, Vtddb, and Vox-rupture representthe gate-body voltage, the time-dependent dielectric breakdown voltageof the gate, and the gate oxide rupture voltage, respectively. When thetransistor (T1) is in OFF state, DC voltage (Vdd2) is applied to theN-well (NW) of transistor (T1). DC voltage (Vdd2) may be chosen suchthat when transistor (T1) is in the OFF state, the voltage across theN-well-body junction stays within a tolerable range to reduce possibleoverstressing of such a junction during this state. When transistor (T1)is in the ON state, DC voltage (Vdd1) is applied to the N-well oftransistor (T1). The person skilled in the art will understand thatswitching of the N-well can improve linearity, consistently with thediscussion above. Additionally, if the body bias and the well bias areswitched at the same time and by the same amount, the body-to-wellvoltage will not change, with the consequence that there is no chargingor discharging current of the body to N-well junction capacitance andassociated time constants. The same techniques may be used for a PMOSdevice. In such case, the polarities of the bias signals and the dopingof regions shown in FIG. 2A may be reversed.

For the sake of added clarification, an exemplary case is consideredwhere DC voltages (V11, V12, V21, V22) are chosen as V11=+3.4V, V12=−3V,V21=0V and V22=−3V. In this case, when transistor (T1) is ON, the gatevoltage is 3.4V and the body voltage is 0V, resulting in a typicallytolerable voltage of 3.4V across the gate-body region. On the otherhand, when the transistor is OFF, Vd=Vs=0V and Vb=−3V, so a reversevoltage of −3V appears across the drain-body and source-body junctionsto ensure a smaller junction capacitance (see also eqs. (1) and (2)),and therefore an improved overall linearity performance of the circuit.This is made possible by virtue of applying proper level shifted biasvoltages to various terminals of transistor (T1) during the ON and OFFstates, thus reducing the negative impacts of drain-body and source-bodycapacitances on the overall linearity performance of the circuit, asdescribed previously. On the other hand, the gate-body region oftransistor (T1) is at V12-V22=0V voltage during the OFF state. Anadditional exemplary case is now considered where the switch device isbiased in an ON state with the gate at 3.4 V and the body at −3 V, whilethe drain and source DC voltages are 0 V. The device will be in thetriode region and a channel will form. The gate to channel voltage willbe 3.4 V and the channel to body voltage will be −3 V, which wouldgenerally be acceptable for a CMOS device with these gate biasconditions. However, if the body is held at −3 V continuously, there maybe transient conditions where the gate and body bias voltages areapplied but the channel has not yet formed. This would result in3.4−(−3)=6.4 V across the gate oxide, which would likely be anoverstress condition for the oxide. In other words, the four voltages(V11, V12, V21, V22) are selected to attain the combined goal of 1)increasing the reverse bias voltage of the drain-body and source-bodyjunctions above the minimum or threshold reverse bias voltage and belowbreakdown voltage in the OFF state of transistor (T1) and 2) maintainingthe voltage across the gate-body junction below set stress voltages inthe ON state of transistor (T1), these stress voltages being Vtddb andVox-rupture as defined previously. In other words, and to reiterate, inthe disabled mode, the reverse bias voltages across source-body anddrain-body (Vsb and Vdb) are maximized while maintaining Vgb<Vtddb andVgb<Vox-rupture in the enabled mode.

In what follows, some exemplary circuits that can benefit from theabove-disclosed teachings are presented.

FIG. 4A shows an exemplary RF switch (400A) comprising a shunt FET stack(420A) and a series FET stack (430A). Each FET stack may include aplurality of transistors, e.g. three, as shown in the figure. When theRF switch (400A) is in the ON state, the transistors within series FETstack (430A) are ON and input signal (440A) is passed from input port(IN) to output port (OUT). In addition, all the transistors within shuntFET stack (420A) are OFF. As a result, input RF signal (440A) isdistributed across non-linear drain-body and source-body capacitances ofthe transistors in shunt FET stack (420A), as shown by signals (450A).Due to the presence of non-linear capacitances across the drain-body andsource-body terminals of the transistors (420A), the presence of suchdistributed signals (450A) may result in an undesired distortion thusdegrading the overall linearity performance of switch (400A).

FIG. 4B shows a further exemplary RF switch (400B) including a shunt FETstack (420B) and a series FET stack (430B). Each FET stack may include aplurality of transistors. When the RF switch (400B) is in an OFF state,the transistors within series FET stack (430B) are OFF, thus decouplinginput port (IN) from output port (OUT). As a result, input RF signal(440B) is distributed across non-linear drain-body and source-bodycapacitances of the transistors in series FET stack (430B), as shown bysignals (450B).

FIG. 4C shows yet another exemplary RF switch (400C) including aplurality of shunt FET stacks (421, . . . , 42 m) and a plurality ofseries FET stacks (431, . . . , 43 m). Each of the mentioned FET stacksmay include a plurality of transistors. RF switch (400C) furtherincludes a plurality of input ports (IN1, . . . , INm) and output port(OUT). As an example, in order to connect input port (IN1) to outputport (OUT), series FET stack (431) is turned ON and the rest of seriesstacks are turned OFF to isolate their corresponding input ports fromoutput port (OUT). Additionally, shunt FET stack (421) is turned OFF andthe rest of shunt FET stacks are ON to short their corresponding inputsignals to ground, thereby providing a better isolation of all inputports other than input port (IN1) from output port (OUT). As a result ofsuch configuration, similarly to what was previously described withregards to switch (400A) of FIG. 4A, input RF signal (441) isdistributed across non-linear drain-body and source-body capacitances ofthe transistors in shunt FET stack (421). This results in undesireddistortion thus degrading the overall linearity performance of switch(400C). Those skilled in the art will understand that these switches arebi-directional and thus assignment of inputs and outputs can be swapped.

The person skilled in the art will also understand that the RF switchesshown in FIGS. 4A-4C are exemplary switches demonstrating some of theapplications that could benefit from the teachings of the presentdisclosure. The teachings presently disclosed can also apply to anyother RF switches and/or circuits implementing devices that are builtthrough a bulk CMOS process and requiring a solution to non-linearcapacitance problems as described above.

In order to overcome the problem shown in FIGS. 4A-4C, FIG. 5 shows anRF switch (500) in accordance with embodiments of the presentdisclosure. RF switch (500) comprises a shunt FET stack (520) and aseries FET stack (530). Each FET stack may include a plurality oftransistors. RF switch (500) further comprises control circuits (510,510′) receiving control signals (Vc, Vc′). Control circuit (510)comprises level shifters (L1, L2, L3). Each of level shifters (L1, L2)may be configured to provide DC voltages (V11, V12), and (V21, V22) attheir respective outputs. Level shifter (L3) may be configured toprovide DC voltages (VDD1, VDD2) at level shifter (L3) output (V3). Withreference to FIG. 5 , all the nodes with labels V3 are coupled to theoutput of level shifter (L3). Control circuit (510′) comprises levelshifters (L1′, L2′, L3′). Each of level shifters (L1′, L2′) may beconfigured to provide DC voltages (V11′, V12′), and (V21′ V22,′) attheir respective outputs. Level shifter (L3′) may be configured toprovide DC voltages (VDD1′, VDD2′) at level shifter (L3′) output (V3′).With continued reference to FIG. 5 , within shunt FET stack (520), eachshunt gate resistor (Rg) is connected to the gate terminal of acorresponding shunt transistor at one end, and to the output of levelshifter (L1) at the other end. Additionally, each shunt body resistor(Rb) is connected to the body terminal of a corresponding shunttransistor at one end, and to the output of level shifter (L2) atanother end. Moreover, each shunt N-well resistor (Rnw) is connected toan N-well of a corresponding shunt transistor at one end and to levelshifter (L3) output (V3) at another end. Similarly, within series FETstack (530), each series gate resistor (Rg) is connected to the gateterminal of a corresponding series transistor at one end, and to theoutput of level shifter (L1′) at the other end. Additionally, eachseries body resistor (Rb) is connected to the body terminal of acorresponding series transistor at one end, and to the output of levelshifter (L2′) at another end. Moreover, each series N-well resistor(Rnw) is connected to an N-well of a corresponding series transistor atone end and to level shifter (L3′) output (V3′) at another end.

With combined reference to FIGS. 3 and 5 , the teachings with regards totransistor (T1) of FIG. 3 and the way such transistor is biased usingcontrol circuit (310) apply to each of the transistors within the shuntstack (520) and their interaction with control circuit (510). The sameapplies to each of the transistors within the series stack (530) andtheir interaction with control circuit (510′). As an example, if atransistor within the shunt FET stack (520) is considered, DC voltages(V11, V12, V21, V22, VDD1, VDD2) feeding level shifters (L1, L2, L3) arechosen such that:

-   -   when the transistor is in an OFF state, the gate voltage, the        body voltage and the well voltage of the transistor are selected        to apply a high enough reverse voltage (e.g. −3V) across        drain-body, source-body and well-body junctions of the        transistor while maintaining the transistor in the OFF state, to        minimize the non-linear capacitances of such junctions and        improve the overall linearity of the circuit; and    -   when the transistor in an ON state, the gate voltage and the        body voltage of the transistor are selected to apply a tolerable        DC voltage (e.g. around 3.5V) across the gate-body junction of        the same transistor in order to avoid overstressing of such a        transistor.        The same applies to each of the transistors within the series        FET stack (530) and their corresponding level shifters (L1′,        L2′, L3′) and DC voltages (V11′, V12′, V21′, V22′, VDD1, VDD2).

With continued reference to FIG. 5 , in accordance with variousembodiments of the present disclosure:

-   -   each of the DC voltages (V11, V12, V21, V22, VDD1, VDD2) may be        equal or different from their (V11′, V12′, V21′, V22′, VDD1′,        VDD2′) counterparts. As an example, in the same circuit, V11 may        be equal to V11′, V21 may be different from V21′, and VDD1 may        be different from VDD1′,    -   each resistor shown with a label, may have a resistance equal to        or different from another resistor shown with the same label.        For example, the gate resistor (Rg) of a transistor within the        shunt FET stack (520) may have a resistance which may be equal        to or different from the resistance of another gate resistor        within the same shunt series stack (520) or within the series        FET stack (530),    -   any of the shunt FET stack (520) and series FET stack (530) may        include one or more transistors,    -   the number of transistors in the shunt FET stack (520) may be        equal to or different from the number of transistors within the        series FET stack (530).

With further reference to FIG. 5 , the person skilled in the art willunderstand that the teachings disclosed with reference to the exemplarycircuit (500) will also apply to any multiple-pole multiple throw switchwith an arbitrary number of shunt FET stacks and series FET stacks, eachFET stack including an arbitrary number of stacked transistors.

With reference to FIGS. 3 and 5 , according to the teachings of thepresent disclosure, control circuits (310, 510, 510′) in conjunctionwith their corresponding shunt or series switches, may be implemented asa part of an integrated circuit which can also be part of a moduleand/or a communication system or device.

A number of embodiments of the invention have been described. It is tobe understood that various modifications may be made without departingfrom the spirit and scope of the invention. For example, some of thesteps described above may be order independent, and thus can beperformed in an order different from that described. Further, some ofthe steps described above may be optional. Various activities describedwith respect to the methods identified above can be executed inrepetitive, serial, or parallel fashion.

It is to be understood that the foregoing description is intended toillustrate and not to limit the scope of the invention, which is definedby the scope of the following claims, and that other embodiments arewithin the scope of the claims. (Note that the parenthetical labels forclaim elements are for ease of referring to such elements, and do not inthemselves indicate a particular required ordering or enumeration ofelements; further, such labels may be reused in dependent claims asreferences to additional elements without being regarded as starting aconflicting labeling sequence).

As should be readily apparent to one of ordinary skill in the art,various embodiments of the invention can be implemented to meet a widevariety of specifications. Unless otherwise noted above, selection ofsuitable component values is a matter of design choice and variousembodiments of the invention may be implemented in any suitable ICtechnology (including but not limited to MOSFET structures), or inhybrid or discrete circuit forms. Integrated circuit embodiments may befabricated using any suitable substrates and processes, including butnot limited to standard bulk silicon, silicon-on-insulator (SOI), andsilicon-on-sapphire (SOS). Unless otherwise noted above, the inventionmay be implemented in other transistor technologies such as bipolar,GaAs HBT, GaN HEMT, GaAs pHEMT, and MESFET technologies. However, theinventive concepts described above are particularly useful with anSOI-based fabrication process (including SOS), and with fabricationprocesses having similar characteristics. Fabrication in CMOS on SOI orSOS processes enables circuits with low power consumption, the abilityto withstand high power signals during operation due to FET stacking,good linearity, and high frequency operation (i.e., radio frequencies upto and exceeding 100 GHz). Monolithic IC implementation is particularlyuseful since parasitic capacitances generally can be kept low (or at aminimum, kept uniform across all units, permitting them to becompensated) by careful design.

Voltage levels may be adjusted or voltage and/or logic signal polaritiesreversed depending on a particular specification and/or implementingtechnology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletionmode transistor devices). Component voltage, current, and power handlingcapabilities may be adapted as needed, for example, by adjusting devicesizes, serially “stacking” components (particularly FETs) to withstandgreater voltages, and/or using multiple components in parallel to handlegreater currents. Additional circuit components may be added to enhancethe capabilities of the disclosed circuits and/or to provide additionalfunctionality without significantly altering the functionality of thedisclosed circuits.

The invention claimed is:
 1. A radio frequency (RF) switch comprising afield effect transistor (FET), the FET comprising a first well and asecond well, wherein: the second well is of an opposite semiconductorpolarity to the first well and a substrate of the FET; the FET isfabricated with a bulk complementary metal-oxide-semiconductor (CMOS)process and is configured to i) receive a gate bias voltage switchablebetween a gate first bias voltage level and a gate second bias voltagelevel to put the FET in an ON or OFF state, respectively; and ii)receive a first well bias voltage switchable between a first well firstbias voltage level in correspondence with the ON state, and a first wellsecond bias voltage level in correspondence with the OFF state; iii)receive a second well bias voltage switchable between a second wellfirst bias voltage level in correspondence with the ON state, and asecond well second bias voltage level in correspondence with the OFFstate, and a first well bias voltage difference between the first wellfirst bias voltage level and the first well second bias voltage level isthe same as a second well bias voltage difference between the secondwell first bias voltage level and the second well second bias voltagelevel; when transitioning from one state to another, the first and thesecond well bias voltages are configured to switch at the same time, andthe gate second bias voltage is negative with respect to ground.
 2. TheRF switch of claim 1, further configured to: receive a first well biasvoltage switchable between a first well first bias voltage level incorrespondence with the ON state, and a first well second bias voltagelevel in correspondence with the OFF state, and receive a second wellbias voltage switchable between a second well first bias voltage levelin correspondence with the ON state, and a second well second biasvoltage level in correspondence with the OFF state; and wherein a secondbody bias voltage level is negative with respect to ground.
 3. The RFswitch of claim 2, wherein the FET comprises a first well to second welljunction, reverse biased in a steady state condition.
 4. The RF switchof claim 3, wherein the FET is an NMOSFET, the first well is p-type, thesecond well is n-type and the substrate is p-type.
 5. A circuitalarrangement comprising the RF switch of claim 2 and a control circuitconfigured to provide the second well bias voltage.
 6. A circuitalarrangement comprising: the RF switch of claim 3; and a control circuitconfigured to provide the gate bias voltage, the first well bias voltageand the second well bias voltage to the FET.
 7. The circuitalarrangement of claim 6, wherein the control circuit comprises aplurality of level shifters configured to provide respective DC gate,first well and second well bias voltages to the FET.
 8. The circuitalarrangement of claim 7, wherein the plurality of level shifters comprisethree level shifters.
 9. The circuital arrangement of claim 8, furthercomprising series gate, first well and second well resistors between theFET and respective level shifters of the plurality of level shifters.10. An electronic module comprising the RF switch of claim
 1. 11. Acommunication device comprising the RF switch of claim
 1. 12. The RFswitch of claim 1, wherein the first gate bias voltage level is +3.4 Vand the second gate bias voltage level is −3 V, the first well firstbias voltage level is 0 V and the first well second bias voltage levelis −3 V.
 13. The RF switch of claim 1, wherein the first gate biasvoltage level, the second gate bias voltage level, the first well firstbias voltage level and the first well second bias voltage level areconfigured to be selected to i) maintain a voltage across thedrain-first well and source-first well junctions of the FET below adiode forward voltage and below a breakdown voltage in the OFF state ofthe FET; and ii) maintain voltage across a gate-first well junction ofthe FET in the ON state of the FET below a gate breakdown voltage and agate oxide rupture voltage of the FET.
 14. A method of biasing a radiofrequency (RF) switch comprising a field effect transistor (FET) switch,the FET switch manufactured using a bulk CMOS process, the methodcomprising: in an ON state of the FET switch: applying a first levelshifted gate voltage to a gate terminal of the FET switch, applying afirst level shifted first well voltage to a first well terminal of theFET switch, and applying a first level shifted second well voltage to asecond well terminal of the FET switch; in an OFF state of the FETswitch: applying a second level shifted gate voltage different from thefirst gate voltage to the gate terminal of the FET switch; applying asecond level shifted first well voltage different from the first wellvoltage to the first well terminal of the FET switch, and applying, atthe same time of the second level shifted first well voltage, a secondlevel shifted well voltage different from the first well voltage to thesecond well terminal of the FET switch, wherein the second level shiftedgate voltage and the second level shifted first well voltage are bothnegative with respect to ground, and wherein a first well voltagedifference between the first level shifted first well voltage and thesecond level shifted first well voltage is the same as a second wellvoltage difference between the first level shifted second well voltageand the second level shifted second well voltage.
 15. The method ofclaim 14, wherein the first level shifted gate voltage, the second levelshifted gate voltage, the first level shifted first well voltage and thesecond level shifted first well voltage are selected to i) maintain avoltage across the drain-first well and source-first well junctions ofthe FET switch below a diode forward voltage and below a breakdownvoltage in the OFF state of the FET switch; and ii) maintain voltageacross a gate-first well junction of the FET switch in the ON state ofthe FET switch below a gate breakdown voltage and a gate oxide rupturevoltage of the FET switch.